Self-calibrated power amplifier linearizers

ABSTRACT

An amplifier linearizer includes a signal adjuster having an internal signal, and an adaptation controller for monitoring the signal adjuster. The internal signal at an input to the adaptation controller is deemed a monitor signal. The adaptation controller generates a control signal for the signal adjuster by accounting for a difference between the internal and monitor signals.

This application is a division of application Ser. No. 09/982,628 filedon Oct. 18, 2001 now U.S. Pat. No. 6,734,731, which claims priority toU.S. Patent Application No. 60/301,978, filed on Jun. 28, 2001, theentire disclosure of each application being incorporated herein byreference.

FIELD OF THE INVENTION

This application generally pertains to, but is not limited to,linearizers used in power amplifiers, for example, RF power amplifiersused in wireless communication systems.

BACKGROUND OF THE INVENTION

RF power amplifiers, like most amplifiers, are substantially linear atsmall signal amplitudes. However, it is preferable to drive poweramplifiers near saturation to deliver significant output power at areasonable efficiency. As the operation of a power amplifier approachessaturation, it will become more nonlinear, and thus, exhibit moredistortion in its output. Consequently, numerous “linearizer” circuitshave been developed over the years in an attempt to remove the poweramplifier's nonlinearity and thereby reduce the distortion in itsoutput. Because the characteristics of the power amplifier may changeover time and frequency, these linearizer circuits may be designed toadapt to present amplifier conditions. A generic power amplifierlinearizer is shown in FIG. 1, and uses either predistortion circuitry,feedforward circuitry, or a combination of both, to correct for thepower amplifier's nonlinearity. (The inclusion of this genericlinearizer in this Background Section is not intended to imply that thecircuit configuration shown therein, and variations thereof, are in theprior art.)

For example, a linearizer may use only a predistortion adjuster circuitp. As will be appreciated by those skilled in the art, in thislinearizer the signal adjuster circuit s is merely a delay line ideallymatching the total delay of the adjuster circuit p and the poweramplifier. In this case, the distortion cancellation circuit, comprisingthe distortion adjuster circuit d, the error amplifier and the delaycircuit, is not used—the output of the linearizer is the output of thesignal power amplifier. The goal of the adjuster circuit p is topredistort the power amplifier input signal so that the power amplifieroutput signal is proportional to the input signal of the linearizer.That is, the predistorter acts as a filter having a transfercharacteristic which is the inverse of that of the power amplifier,except for a complex constant (i.e., a constant gain and phase). Becauseof their serial configuration, the resultant transfer characteristic ofthe predistorter and the power amplifier is, ideally, a constant gainand phase that depends on neither frequency nor signal level.Consequently, the output signal will be the input signal amplified bythe constant gain and out of phase by a constant amount, that is,linear. Therefore, to implement such predistortion linearizers, thetransfer characteristic of the power amplifier is computed and apredistortion filter having the inverse of that transfer characteristicis constructed. Preferably, the predistortion filter should alsocompensate for changes in the transfer function of the power amplifier,such as those caused by degraded power amplifier components.

Other linearizers use feedforward circuitry to correct for thenonlinearity in the power amplifier. A feedforward linearizer usuallyuses a combination of signal adjuster circuit s 110 and distortionadjuster circuit d 111 as configured in FIG. 1 (in this linearizer,predistortion adjuster p 109 is not used). In an alternativeconfiguration, the signal adjuster circuit may be placed before thepower amplifier, i.e., as adjuster circuit p 109, an example of which isshown in FIG. 7. This latter configuration advantageously compensatesfor any additional signal distortion caused by the signal adjustercircuit, since it will be superimposed upon the distortion caused by thepower amplifier and be removed by a distortion cancellation circuit.Further, adjusters p 109, s 110, and d 111 may be used simultaneously tolinearize the power amplifier.

As shown in FIG. 1, a feedforward linearizer comprises two maincircuits: a signal cancellation circuit 101 and a distortioncancellation circuit 102. The RF signal is input to the signal poweramplifier 103, which as discussed above, is assumed to be operating in anon-linear range and thus distorting the output signal. The signalcancellation circuit 101 ideally subtracts a linear estimate of the RFsignal from the distorted power amplifier output signal so that only thenonlinear distortion signal (or “error signal”) (v_(e)) remains. As willbe appreciated to those skilled in the art, the signal pickoff points,the adder 104, and the subtractors 106 and 107 shown in FIG. 1 and otherfigures may be implemented by directional couplers, splitters orcombiners, as appropriate. In the distortion cancellation circuit 102,the distortion signal is adjusted and amplified by error amplifier 108to match the distortion signal component of the power amplifier outputsignal delayed by delay 112. The amplified distortion signal is thensubtracted from the output of delay 112 by subtractor 107 to provide thelinearizer output signal v_(o). The linearizer output signal is asubstantially distortion-free amplified RF signal, the output that wouldhave been obtained if the power amplifier were truly linear.

Generally, the adjuster circuits discussed above do not necessarily allhave the same structure—adjuster circuits p 109, s 110 and d 111 may allbe implemented with different circuitry. For example, the adjustercircuit p 109 may be a nonlinear polynomial filter, while the adjustercircuits s 110 and d 111 may be finite impulse response (FIR) filters.In addition, some methods of controlling these adjuster circuits mayemploy pilot (tone) signals generated by an optional pilot signalgenerator 113.

The relationship of the input and output signals of an adjuster circuitdepends on the settings of one or more parameters of that adjustercircuit, as will be discussed in further detail below. Duringadaptation, the values of one or more internal signals of an adjustercircuit are used to determine appropriate settings for its parameters.As shown in FIG. 1, an “adaptation controller” 114 monitors the errorand output signals v_(e) and v_(o), and in some cases, the internaladjuster signals. (In FIG. 1 and other figures, a stroke on an arrowdenotes a multiplicity of signals or a multiplicity of parameters, asthe case may be.) On the basis of the monitored signal values, and inaccordance with the adaptation algorithm, the adaptation controller setsthe adjuster circuit parameters.

For example, a three-branch adaptive polynomial predistortion adjustercircuit p 109 is shown in FIG. 2. The upper branch 200 is linear, whilethe middle branch has a nonlinear cubic polynomial filter 201 and thelower branch has a nonlinear quintic polynomial filter 202, theimplementation of which nonlinear filters is well known to those skilledin the art. Each branch also has a complex gain adjuster (“CGA”),respectively 203, 204, and 205, to adjust the amplitude and phase of thesignal as it passes therethrough. By setting the parameters (GA, GB) ofeach of the CGAs, a polynomial relationship between the input and outputof the adjuster circuit can be established to compensate for amemoryless nonlinearity in the power amplifier. The adaptationcontroller, via a known adaptation algorithm, uses the input signal, theoutput of the nonlinear cubic polynomial filter, the output of thenonlinear quintic polynomial filter, and the error signal (the poweramplifier output signal minus an appropriately delayed version of inputsignal) to generate the parameters (GA, GB) for the three CGAs.Generally, the adaptation algorithm is selected to minimize a certainparameter related to the error signal (for example, its power over apredetermined time interval). Examples of such adaptation algorithms aredescribed in more detail below.

Two possible CGA implementations are respectively shown in FIGS. 3A and3B. The implementation shown in FIG. 3A uses polar control parameters GAand GB, where GA sets the amplitude of the attenuator 301, while GB setsthe phase of the phase shifter 302. The implementation shown in FIG. 3Buses Cartesian control parameters, also designated GA and GB, where GAsets the real part of the complex gain, while GB sets the imaginary partof the complex gain. In this implementation, the input signal I is splitinto two signals by splitter 306, one of which is then phase-shifted by90 degrees by phase shifter 303, while the other is not. After Ga. andGB are applied by mixers or attenuators 305 and 304 respectively, thesignals are summed by combiner 307 to produce the CGA output signal O.U.S. Pat. No. 6,208,207 describes, in part, the linearization of thesemixers and attenuators, so that desired values of complex gain can bepredictably obtained by appropriate setting of the control voltages GAand GB.

For example, a three-branch adjuster circuit s 110 and a three-branchadjuster circuit d 111 in a feedforward linearizer are shown in FIG. 4.Feedforward linearizers having one or more branches in the adjustercircuits are described in U.S. Pat. Nos. 5,489,875 and 6,208,207, bothof which are incorporated herein by reference. Each branch within thecircuits 110 and 111, labeled “FIR adjuster”, includes a delay element(i.e., delays 401, 403 and 405 in the adjuster s 110, and delays 407,409, and 411 in adjuster d 111) and a CGA (i.e., CGAs 402, 404 and 406in the adjuster s 110, and CGAs 408, 410, and 412 in adjuster d 111).The delays in each branch may be different, and the sum of the parallelbranches act as an analog FIR filter (also known as an analogtransversal filter).

Appropriate settings of the parameters (GA, GB) of the CGAs allow thefirst FIR adjuster circuit 110 to mimic the linear portion of the poweramplifier response, including the effects of amplifier delay and otherfiltering, and for frequency dependence of its own components. Ideally,the amplifier nonlinear distortion is revealed at the output of thesubtractor following the first FIR adjuster circuit (v_(e)). Appropriatesettings of the parameters (GA, GB) of the second FIR adjuster circuit111 allow it to compensate for delay and other filtering effects in theamplifier output path and for frequency dependence in its owncomponents, and to subtract a replica of the nonlinear distortion fromthe delayed amplifier output. The adaptation controller 114 of FIG. 4,via a known adaptation algorithm, uses the internal signals of thebranches of the signal and distortion adjuster circuits s 110 and d 11and their respective error signals v_(e) and v_(o), to compute GA and GBfor each of the CGAs in the signal and distortion adjuster circuits 110and 111. In this fashion, the linearization circuit compensates for anamplifier nonlinearity with memory. Examples of such adaptationcontrollers can be found in U.S. Pat. Nos. 5,489,875 and 6,208,207.

The linearizer circuits of the prior art, however, ignore a phenomenonthat often determines the success or failure of the adaptationcontroller—the monitored signals, as measured by the adaptationcontroller, are not necessarily equal to their counterpart internalsignals within the adjuster circuits, or to the actual error and outputsignals v_(e) and v_(o), as the case may be. The reason is that thecables, circuit board traces, and other components in the signal pathsthat convey the internal adjuster circuit signals, or the error andoutput signals v_(e) and v_(o), to the adaptation controller introduceinadvertent phase and amplitude changes into those signals. The truesituation is represented in FIGS. 6 and 8, where these phase andamplitude changes are generically modeled as “observation filters”(601-603, 804, 805). In addition, H_(p)(f) 601, H_(s)(f) 602 andH_(d)(f) 603 may each be considered to comprise a bank of “observationsubfilters,” such as h_(p1)(f), h_(p2)(f), etc., each observationsubfilter modeling the transformation of a particular internal signal ofan adjuster circuit into a corresponding monitor signal (for example,one observation subfilter per branch of a multibranch adjuster). Thecharacteristics of these observation filters or subfilters are initiallyunknown.

In the simplest case, the observation filters or subfilters mayrepresent fixed amplitude and phase changes on each of the signal paths.In a more complex case, however, the amplitude and phase changes, andthus the observation filters, can be frequency-dependent. For example, athree-branch signal adjuster p 109 located in front of the poweramplifier 103 is shown in FIG. 7. This adjuster circuit is constructedso that each branch k (k=0, 1, 2) contains a frequency-dependent filterg_(k)(f) (701, 703 or 705), which serves as a generalization of thedelay elements of an FIR adjuster, and a CGA (702, 704 or 706). (Themention of these general branch filters g_(k)(f) in this BackgroundSection is not intended to imply that their use in FIR adjuster circuitsis known in the prior art; rather, such use is intended to be within thescope of the present invention.) Each observation subfilter h_(k)(f)(707, 708 or 709) of observation filter H_(p)(f) 601 models thetransformation of the internal signal on branch k into the correspondingmonitor signal used by the adaptation controller.

It should be understood that placement of filters (or filter banks) asshown in FIG. 6 is just one of many ways to model the difference(inequality) between the internal signals and the monitor signals.Nevertheless, the observation filters shown in FIG. 6 are sufficient,because other ways of modeling the difference between internal signalsand their corresponding monitor signals are equivalent to therepresentation thereto. For example, FIG. 8 shows a linearizer circuitthat includes an observation filter h_(em)(f) 804 in the path of theerror signal v_(e) output from the first subtractor 106, and anobservation filter h_(om)(f) 805 in the path of the RF output signalv_(o), output from the second subtractor 107, to the adaptationcontroller 114. These observation filters can be transformed to therepresentation shown in FIG. 6 by including the effect of h_(em)(f) 804in the branch paths of adjuster circuits p 109 and s 110, and ofh_(om)(f) 805 into the branch path of adjuster circuit d 111 and thedistortion cancellation circuit reference branch.

The severity of the problem caused by the differences between theinternal adjuster signals and their corresponding monitor signals can beillustrated by a simple example. FIG. 9 illustrates the signalcancellation circuit 101 of a single-branch feedforward linearizer.Specifically, the signal adjuster circuit s 110 includes a single delay901 followed by a CGA 902. The adaptation controller 114 uses a known“stochastic gradient” algorithm (see, for example, the gradientadaptation controller disclosed in U.S. Pat. No. 5,489,875) to correlateusing bandpass correlator 903 the error signal at the output of thesubtractor with the monitored replica of the internal signal of theadjuster circuit, both of which are bandpass signals. The controllerintegrates the result using integrator 905, via loop gain amplifier 904,to provide CGA parameters GA and GB. The internal structure of a knownbandpass correlator is depicted in FIG. 10, and includes a phase shifter1001, mixers 1002 and 1003, and bandpass filters (or integrators) 1004and 1005 (for a description of the operation of such a bandpasscorrelator, see FIG. 3 of U.S. Pat. No. 5,489,875 and the textcorresponding thereto). In the idealized situation considered in theprior art, the monitor signal and the internal signal are equal, withH_(s)(f)=h=1 (906) at all frequencies, and the correlation result is astochastic estimate of the gradient of the error signal power withrespect to the CGA parameters. That is, the correlation result isproportional to the change in CGA parameter settings that would resultin the greatest increase in error signal power. Sign reversal andintegration causes the CGA parameters to be corrected in the directionthat most decreases the error signal power, and the adaptation loopconverges correctly to the setting of GA and GB that minimizes errorsignal power, with a time constant determined by the value K of the loopgain amplifier 904.

To continue this example, when the linearizer is implemented, it willdiffer from the ideal case in that the monitor signal (bandpass signal 1of FIG. 10) will likely have unknown phase and amplitude shifts withrespect to the internal adjuster signal. These shifts are represented bythe complex variable h (906) in FIG. 9. If h has a phase shift of 180degrees, then the correlation result will be negated, and the correctionto the CGA parameters will be made in a direction that maximallyincreases, rather than decreases, the error signal power. This, in turn,will cause the circuit to diverge from its ideal setting. Moregenerally, a phase shift in h of over 90 degrees will cause the circuitto diverge. A phase shift value of greater than zero degrees, but lessthan 90 degrees, will allow the circuit to converge to its idealsetting, but with decreasing rapidity as it approaches 90 degrees.

The same problem may afflict adaptation controllers based on otheralgorithms that exploit the relationships among monitored signals inorder to make corrections to CGA settings. For example, a least squares(“LS”) algorithm or a recursive LS algorithm may also diverge (orconverge more slowly) under the same phase shift conditions as set forthabove for the stochastic gradient algorithm.

In a multibranch adjuster circuit, for example, the polynomialpredistorter circuit of FIG. 2 or the feedforward circuit of FIGS. 4 and6, there are further consequences of the lack of equality between aninternal adjuster signal and its corresponding monitor signal. Signalscarried on the branches of an adjuster tend to be highly correlated,making stochastic gradient adaptation slow. The remedy is lineartransformation of the multiple branch signals to produce a multiplicityof decorrelated signals. The decorrelated signals, or modes, are thenadapted individually to provide a much faster convergence. However, thelack of equality between internal and corresponding monitor signals, asmodeled by the filter banks in FIG. 6, reduces the ability todecorrelate those signals completely. This results in branch signalswith residual correlation, thus reducing the benefits of decorrelation.Furthermore, if the internal and monitor signals are not equal,unfavorable phase and amplitude relationships among the filters maycause one or more of the decorrelation mode adaptations to diverge,preventing adaptation altogether.

In addition, in a stochastic gradient controller, if wide powerdisparities exist among the decorrelated signals, stronger signals mayinterfere or “mask” the weaker signals, degrading the latter and slowingadaptation. To reduce this masking problem, a “partial gradient”algorithm may be used by the adaptation controller (see, for example,the partial gradient adaptation controller disclosed in U.S. Pat. No.5,489,875), in which the correlation between two bandpass signals isapproximated as a sum of partial correlations taken over limitedbandwidths at selected frequencies. By making the frequenciesselectable, correlations may be calculated at frequencies that do notcontain strong signals, so that the strong signals do not mask the weaksignals. In addition, a digital signal processor (DSP) may be used toperform correlation, because the correlations are taken over limitedbandwidths. This eliminates the DC offset that otherwise appears in theoutput of a correlator implemented by directly mixing two bandpasssignals.

FIG. 11 illustrates a partial correlator, in which local oscillators1101 and 1102 select the frequency of the partial correlation. Frequencyshifting and bandpass filtering are performed by the mixer/bandpassfilter combinations 1103/1107, 1104/1108, 1105/1109, and 1106/1110. Thesignals output by the bandpass filters 1109 and 1110 are digitallyconverted, respectively, by A/D converters 1111 and 1112. Those digitalsignals are bandpass correlated by DSP 1113 to produce the real andimaginary components of the partial correlation. (See, for example, FIG.9 of U.S. Pat. No. 5,489,875 for a description of the operation of apartial correlator similar to that shown in FIG. 11 herein.) However, asin the case of the stochastic gradient adaptation controller, a lack ofequality between the internal signals and their corresponding monitorsignals (for example, bandpass signal 1) may cause either divergence orslowed convergence of the partial gradient adaptation controller of FIG.11.

Accordingly, self-calibrated power amplifier linearizers are desired tocompensate for the lack of equality between the internal adjustersignals and their corresponding monitor signals, and to overcome theresulting divergence, or slowed convergence, of the adaptationcontrollers used therein.

SUMMARY OF THE INVENTION

In one aspect of the presented invention, an amplifier linearizerincludes a signal adjuster having an internal signal, and an adaptationcontroller for monitoring the signal adjuster. The internal signal at aninput to the adaptation controller is deemed a monitor signal. Theadaptation controller generates a control signal for the signal adjusterby accounting for a difference between the internal and monitor signals.

This and other aspects of the invention may be ascertained from thedetailed description of the preferred embodiments set forth below, takenin conjunction with the one or more of the following drawings.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram of a generic power amplifier linearizercircuit.

FIG. 2 shows an example of a polynomial predistortion linearizercircuit.

FIGS. 3A and 3B respectively show two configurations of a complex gainadjuster.

FIG. 4 shows an example of a multibranch feedforward linearizer circuit.

FIG. 5 shows a hybrid predistorter and feedforward linearizer circuit.

FIG. 6 is a block diagram of a generic power amplifier linearizercircuit, in which observation filters are included to model the lack ofequality between the internal adjuster signals and the correspondingmonitor signals used by the adaptation controller.

FIG. 7 is a generic FIR signal adjuster circuit showing a bank ofobservation subfilters to model the lack of equality between theinternal signal adjuster circuit signals and the corresponding monitorsignals used by the adaptation controller.

FIG. 8 is a block diagram of a generic power amplifier linearizercircuit, in which observation filters are included to model the lack ofequality between the internal adjuster signals, error output signalv_(e) and RF output signal v_(o), and the corresponding monitor signalsused by the adaptation controller.

FIG. 9 is an example of a one-branch feedforward signal adjustercircuit, and a stochastic gradient adaptation controller using abandpass correlator, in which observation filter h is included to modelthe lack of equality between the internal signal adjuster signal and thecorresponding monitor signal used by the adaptation controller.

FIG. 10 is an example of the bandpass correlator used by the adaptationcontroller shown in FIG. 9.

FIG. 11 is an example of a partial bandpass correlator used in a partialgradient adaptation controller.

FIG. 12 is a two-branch feedforward circuit with observation filtersincluded in the monitor signal paths of the FIR adjuster branches.

FIG. 13 is a two-branch feedforward signal cancellation circuit withobservation filters for the adjuster branches.

FIG. 14 is a multibranch predistorter containing general nonlinearitieswith frequency dependence.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The embodiments of the present invention are directed toself-calibration techniques by which the adaptation controller (1)determines the frequency responses of the observation filters betweenthe internal signals and the monitor signals (and hence, the frequencyresponses of the underlying circuit components modeled by thoseobservation filters) and (2) corrects for these responses, so that themonitored signals are substantially representative of the internalsignals of the adjuster circuits. The benefits of self-calibration are(1) reliable adaptation, without risk of divergence; (2) fasteradaptation than without self-calibration; and (3) access to methods thatdecorrelate the internal signals, explicitly or implicitly, to allowfast adaptation of all the signal modes. This self-calibration can beperformed upon the initialization of the linearizer (for example, atdevice turn-on), or from time-to-time as needed. Further, theself-calibration is performed by the linearizer itself without humanintervention.

FIG. 12 illustrates a feedforward linearizer in which an FIR adjuster s1210 in the signal cancellation circuit is located parallel to the poweramplifier 103 (the case where the FIR adjuster s is in series with thepower amplifier is discussed further below). The FIR adjuster s has twobranches, the upper including a delay 1230 and a CGA 1231, and the lowerincluding a delay 1232 and a CGA 1233. The outputs of the CGAs 1231 and1233 are summed by the combiner 1238. The feedforward linearizer alsohas an FIR adjuster d 1211 in the distortion cancellation circuit. TheFIR adjuster d also has two branches, the upper including a delay 1234and a CGA 1235, and the lower including a delay 1236 and a CGA 1237. Theoutputs of the CGAs 1235 and 1237 are summed by the combiner 1239.Although FIG. 12 illustrates FIR adjusters with two branches, theself-calibration techniques of the present invention apply to adjusterswith one or more branches. In addition, these self-calibrationtechniques are described below with respect to FIR adjuster circuits forsake of clarity. However, those skilled in the art will recognize thatthese techniques apply to a wide range of adjuster circuits havingstructures as described above, including, but without limitation, FIRadjusters, polynomial adjusters, and adjusters with general filters onthe branches.

In the circuit shown in FIG. 12, the complex gains (amplitude and phase)of the observation filters h_(s0)(f) 1220, h_(s1)(f) 1221, h_(e0)(f)1222 and h_(e1)(f) 1223 are first determined. Then those gains are usedto adjust the respective monitor signals v_(am0), v_(am1), v_(bm0) andv_(bm1) so they are representative of the corresponding internaladjuster signals v_(a0), v_(a1), v_(b0) and v_(b1).

In one embodiment, it is assumed that the observation filters do notdepend on frequency, so that they are each characterized by a singlecomplex gain, i.e., h_(s0), h_(s1), h_(e0), and h_(e1) With reference toFIG. 12, the adaptation controller 1214 determines the complex gainh_(s0) as follows:

(1) set the power amplifier 103 in standby mode, so that its output iszero;

(2) set the CGA 1233 complex gain a₁ to zero through an appropriatechoice of the control voltages, so that the corresponding CGA output iszero;

(3) set the CGA 1231 complex gain a₀ to some nominal value a₀′ throughappropriate choice of control voltages;

(4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 to generate atone for calibration;

(5) in the adaptation controller, use a bandpass correlator (forexample, the bandpass correlator shown in FIG. 10) to produce thecorrelation of signal v_(e) with monitor signal v_(am0); the result is:

C _(eam0) =a ₀ ′*h _(s0) ^(*) *P _(a0),

where the asterisk denotes complex conjugation and P_(a0) denotes thepower of signal v_(a0);

(6) in the adaptation controller, use a bandpass correlator to producethe correlation of monitor signal v_(am0) with itself; the result is:

C _(am0) =|h _(s0)|² *P _(a0),

where the bars denote the magnitude of a complex quantity; and

(7) determine the observation filter gain as:

h _(s0) a ₀ ′*C _(am0) /C _(eam0).

The gain h_(s1) is determined in a similar fashion with branch “0” setto zero (a₀=0) and branch “1” enabled (a₁=a₁′).

The adaptation controller 1214 determines the complex gain h_(e0) asfollows:

(1) set the power amplifier in standby mode, so that its output is zero,and set at least one of the CGA gains (a₀ or a₁) in the signal FIRadjuster s to a non-zero value, so that the power of the error signalP_(e) is non-zero;

(2) set the CGA 1237 complex gain b₁ to zero through an appropriatechoice of the control voltages, so that the corresponding CGA output iszero;

(3) set the CGA 1235 complex gain b₀ to some nominal value b₀″ throughappropriate choice of control voltages;

(4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 to generate atone for calibration;

(5) in the adaptation controller, use a bandpass correlator to producethe correlation of signal v_(o) with monitor signal v_(bm0); the resultis:

 C _(obm0) =b ₀ ′*h _(e0) ^(*) *P _(b0),

where the asterisk denotes complex conjugation and P_(b0) denotes thepower of signal v_(b0);

(6) in the adaptation controller, use a bandpass correlator to producethe correlation of monitor signal v_(bm0) with itself; the result is:

C _(bm0) =|h _(e0)|² *P _(b0),

where the bars denote the magnitude of a complex quantity; and

(7) determine the observation filter gain as:

h _(e0) =b ₀ ′*C _(bm0) /C _(obm0).

The gain h_(e1) is determined in a similar fashion with branch “0” setto zero (b₀=0) and branch “1” enabled (b₁=b₁′).

In another embodiment, it is assumed that the observation filters dependon frequency. Consequently, to approximate their frequency responses,the adaptation controller determines their gains at a selected set of Nfrequencies f_(i), i=1, 2, . . . , N. The adaptation controller 1214determines the gain h_(s0)(f₁) at frequency f₁ as follows:

(1) set the power amplifier 103 to standby mode, so that its output iszero;

(2) set the CGA 1233 gain of a₁ to zero through appropriate choice ofthe control voltages, so that the CGA output is zero;

(3) set the CGA 1231 gain a₀ to some nominal value a₀′ throughappropriate choice of control voltages;

(4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 set tofrequency f₁;

(5) use a partial correlator (for example, the partial correlator shownin FIG. 11), with local oscillators set to select frequency f₁, toproduce the correlation of signal v_(e) with monitor signal v_(am0); theresult is:

C _(eam0)(f ₁)=a ₀ *h _(s0)*(f ₁)*P _(a0)(f ₁),

where P_(a0)(f₁) denotes the power of signal v_(a0) at frequency f₁;

(6) use a partial correlator, with local oscillators set to selectfrequency f₁, to produce the correlation of monitor signal v_(am0) withitself; the result is:

C _(am0)(f ₁)=|h _(s0)(f ₁)|² *P _(a0)(f ₁);

(7) determine the observation filter gain at frequency f₁ as:

h _(s0)(f ₁)=a ₀ ′C _(am0)(f ₁)/(C _(eam0)(f ₁)).

Similarly, the adaptation controller 1214 determines the gain h_(e0)(f₁)at frequency f₁ as follows:

(1) set the power amplifier 103 to standby mode, so that its output iszero, and set at least one of the CGA gains (a₀, a₁) in the signalcancellation circuit to a non-zero value, so that the power of the errorsignal P_(e)(f₁) is non-zero;

(2) set CGA 1237 gain b₁ to zero through appropriate choice of thecontrol voltages, so that the corresponding CGA output is zero;

(3) set the CGA 1235 gain b₀ to some nominal value b₀′ throughappropriate choice of control voltages;

(4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 set tofrequency f₁;

(5) use a partial correlator, with local oscillators set to selectfrequency f₁, to produce the correlation of signal v₀ with monitorsignal v_(bm0); the result is:

C _(obm0)(f _(i))=b ₀′*h_(e0)*(f ₁)*P _(b0)(f ₁),

where P_(b0)(f₁), denotes the power of signal v_(b0) at frequency f₁;

(6) use a partial correlator, with local oscillators set to selectfrequency f₁, to produce the correlation of monitor signal v_(bm0) withitself; the result is:

C _(bm0)(f _(j))=|h _(e0)(f ₁)|² *P _(b0)(f ₁);

(7) determine the observation filter gain at frequency f₁ as:

h _(e0)(f ₁)=b ₀ ′C _(bm0)(f ₁)/(C _(obm0)(f ₁))

The complex gains h_(s0)(f_(i)) and h_(e0)(f_(i)) at frequencies i=2, 3,. . . , N are determined similarly. The frequency responses of theremaining observation filters h_(s1)(f) and h_(e1)(f) are determined byselecting them one at a time through choice of CGA gains, and thenrepeating the above-described methods for each frequency f_(i), i=1, 2,. . . , N.

In the above procedures, the frequencies f_(i) at which calibration isobtained are the same for the signal and distortion cancellationcircuits. However, the frequencies f_(i) at which calibration isobtained may differ between the signal and distortion cancellationcircuits.

To self-calibrate a single-branch adjuster circuit, step (2) of theabove-described methods is eliminated. To self-calibrate an adjustercircuit having more than two branches, one branch at a time is enabled,while all others are set to zero, until all corresponding observationfilter gains are determined.

Computing the complex gains of the observation filters for the signalcancellation circuit may be done without computing the same for thedistortion cancellation circuit, and vice-versa. Also, computing thecomplex gains of one or more of the observation filters for thedistortion cancellation circuit may be done prior to computing the samefor the signal cancellation circuit, and vice-versa.

Moreover, although the adjusters in this embodiment had delay lines inthe branches, as shown in FIG. 12, the procedures for estimating theobservation filter complex gains, whether they be frequency-independentor frequency-dependent, are equally applicable to adjusters constructedwith filters of any type in place of one or more of the delay lines1230, 1232, 1234 and 1236.

Once the observation filter complex gains are computed, thecorresponding monitor signals need to be appropriately adjusted. Theadaptation controller divides the monitor signals by the respectiveobservation filter complex gains (either frequency-independent orfrequency-dependent, as the case may be) to approximate the trueinternal adjuster circuit signals. For example, for branch k andfrequency f_(i), the controller calculates:

v _(ak) =v _(amk) /h _(sk)(f _(i)); and

 v _(bk) =v _(bmk) /h _(ek)(f _(i)).

Once these self-calibration procedures are performed, the effect onconvergence is dramatic. This is particularly true when a partialcorrelator is implemented using a DSP. Convergence is reliable androbust in the face of amplitude and phase changes introduced by thecables, circuit board traces, and other components in the signal pathsthat convey the internal signals to the adaptation controller. Moregenerally, any adaptation algorithm, such as stochastic gradient, withor without decorrelation of the branch signals, or least squares, can bemade insensitive to these amplitude and phase changes, because theadaptation controller can always recover the internal signals from themonitor signals by dividing them by the corresponding determinedobservation filter complex gains.

Variations of the self-calibration procedures of the present inventionwill be evident to those skilled in the art. Two are listed here forillustrative purposes.

In one variation (described here only for a two-branch signal adjustercircuit, but equally applicable to a two-branch distortion adjustercircuit, as appropriately modified in accord with the method forcomputing h_(e0)(f_(i)) and h_(e1)(f_(i)) described above), theadaptation controller 1214 simultaneously determines the frequencyresponses of all observation filters at frequency f_(i) as follows:

(1) set the power amplifier 103 to standby mode, so that its output iszero;

(2) set all the CGA gains to non-zero nominal values a_(k)′, k=0, . . .K−1, where k indexes the branch and there are K branches (in FIG. 12,the circuit is illustrated for K=2 branches);

(3) apply an input signal with components at frequency f_(i) to theamplifier, or use an internal pilot signal generator 113 set tofrequency f_(i);

(4) use partial correlators to measure all the pairwise correlationsamong the monitor signals at frequency f_(i); for K=2, this results in:

C _(am0)(f _(i))=P _(a0)(f _(i))*|h _(s0)(f _(i))|²

C _(am1)(f _(i))=P _(a1)(f _(i))*|h _(s1)(f _(i))|²

C _(am01)(f _(i))=P _(a01)(f _(i))*h _(s0)(f _(i))*h _(s1)*(f _(i))

where P_(a0)(f_(i)) denotes the power of v_(a0) at f_(i), P_(a1)(f_(i))denotes the power of v_(a1) at f_(i), and P_(a01)(f_(i)) denotes the“crosspower” (the correlation) of v_(a0) and v_(a1) at f_(i);

(5) use partial correlators to measure the correlation between the errorsignal and each of the monitor signals; for K=2, this results in

C _(eam0)(f _(i))=a ₀ ′*P _(a0)(f _(i))*h _(s0)*(f _(i))+a ₁ ′* P_(a01)*(f _(i))*h _(s0)*(f)

C _(eam1)(f _(i))=a ₀ ′*P _(a01)(f _(i))*h _(s1)*(f _(i))+a ₁ ′* P_(a1)(f _(i))*h _(s1)*(f _(i))

which becomes, after substitution for the powers, a set of equations inthe observation filter gains:

C _(eam0)(f _(i))=a ₀ ′*C _(am0)(f _(i))*h _(s0)(f _(i))⁻¹ +a ₁ ′*C_(am01)*(f _(i))*h _(s1)(f _(i))⁻¹

C _(eam1)(f _(i))=a ₀ ′*C _(am01)(f _(i))*h _(s0)(f _(i))⁻¹ +a ₁ ′*C_(am1)(f _(i))*h _(s1)(f _(i))⁻¹

(6) solve the set of equations for h_(s0)(f_(i))⁻¹ and h_(s1)(f_(i))⁻¹;take their reciprocals to obtain the desired frequency responsesh_(s0)(f_(i)) and h_(s1)(f_(i)).

The method extends in a straightforward way to signal or distortionadjuster circuits with more than two branches.

In a second variation, described here only for the signal adjustercircuit (but equally applicable to distortion adjuster, appropriatelymodified as described above), the adaptation controller 1214 determinesthe responses of the observation filters at frequency f_(i) withoutputting the power amplifier 103 in standby mode. It will be describedhere only for one of the branches (branch k):

(1) set all of the CGA gains to zero;

(2) apply an input signal with components at frequency f_(i) to theamplifier, or use an internal pilot signal generator 113 set tofrequency f_(i); the power of the signal is set to operate the poweramplifier at a nominal operating point;

(3) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(amk); the result is a bias term C_(eamk)′(f_(i));

(4) set the CGA gain a_(k) to some nominal value a_(k)′ throughappropriate choice of control voltages;

(5) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(amk); the result is

C _(eamk)(f _(i))=a _(k) ′h _(sk)*(f _(i))*P_(ak)(f _(i))+C _(eamk)′(f_(i));

(6) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of monitor signal v_(amk);the result is

C _(amk)(f ₁)=|h _(sk)(f _(i))|² *P _(ak)(f _(i));

(7) determine the observation filter gain at frequency f_(i) as

h _(sk)(f _(i))=a _(k) ′*C _(amk)(f _(i))/(C _(eamk)(f _(i))−C′_(eamk)′(f _(i))).

In another aspect of the present invention, the adjuster circuit 1309precedes the power amplifier 103, as shown in FIG. 13, an expandedversion of FIG. 7. The branch filters h_(c0)(f) 1330 and h_(c1)(f) 1332can be as simple as delays or as complex as general linear filters.These filters respectively precede CGAs 1331 and 1332, the outputs ofwhich are summed by combiner 1334. In this model, the amplifier gain isincluded in the branch filter responses. The filter h_(r)(f) 1310 in thereference branch may also be simple or complex; even if such a filter isnot inserted explicitly, h_(r)(f) 1310 represents the response of thebranch. The RF switch 1340 is optional; as explained below, its presenceor absence gives rise to two embodiments. The objective in both cases isto determine the responses of the observation filters h_(p0)(f) 1320 andh_(p1)(f) 1321 at selected frequencies.

In the first of such embodiments, the RF switch 1340 is absent and thereis an unobstructed path from the output of filter h_(r)(f) 1310 to theinput of the subtractor 106. To determine the response h_(pk)(f_(i)) ofthe observation filter k at frequency f_(i) the adaptation controller1314 performs the following actions:

(1) set the power amplifier to standby mode, so that its output is zero;

(2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f_(i); the power of the signal is set to operate the poweramplifier at a nominal operating point;

(3) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(emk); the result is a bias term:

C′ _(ecmk)(f _(i))=−h _(r)(f _(i))*h _(pk)*(f _(i))*h _(ek)*(f _(i))·P_(in)(f _(i)),

where P_(in)(f_(i)) is the input power at frequency f_(i);

(4) restore the power amplifier to operational mode;

(5) set the branch-k CGA gain to some nominal value c′_(k); set allother CGA gains to zero;

(6) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(emk); the result is:

C _(ecmk)(f _(i))=(c′ _(k) *h _(ck)(f _(i))−h _(r)(f _(i)))*h _(pk)*(f_(i))*h _(ck)*(f _(i))*P _(in)(f _(i);)

(7) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of monitor signal v_(cmk)with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² *|h _(ck)(f _(i))|² *P _(in)(f_(i));

(8) determine the branch-k observation filter response at frequencyf_(i) as:

h _(pk)(f _(i))=c′ _(k) C _(emk)(f _(i))/(C _(ecmk)(f _(i))−C′ _(ecmk)(f_(i))).

The observation filter responses for the other filters are determinedsimilarly.

In the second embodiment of this aspect of the present invention, the RFswitch 1340 is present. As will be seen, it simplifies the calibrationprocedure significantly. To determine the response h_(pk)(f_(i)) of theobservation filter k at frequency f_(i), the adaptation controllerperforms the following actions:

(1) open the RF switch 1340, thereby disconnecting the filter h_(r)(f)1310 from the subtractor 106;

(2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f_(i); the power of the signal is set to operate the poweramplifier at a nominal operating point;

(3) set the branch-k CGA gain to some nominal value c′_(k); set allother CGA gains to zero;

(4) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(cmk); the result is:

C _(ecmk)(f _(i))=c′ _(k) |h _(ck)(f _(i))|² h _(pk)*(f _(i))*P _(in)(f_(i)),

where P_(in)(f_(i)) is the input power at frequency f_(i);

(5) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal monitor v_(cmk)with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² *|h _(ck)(f _(i))|² *P _(in)(f_(i));

(6) determine the branch-k observation filter response at frequencyf_(i) as:

h _(pk)(f _(i))=c′ _(k) C _(cmk)(f _(i))/C _(ecmk)(f _(i));

(7) close the RF switch 1340.

The observation filter responses for the other filters are determinedsimilarly.

In still another aspect of the present invention, the adjuster circuit1409 precedes the power amplifier 103, as shown in FIG. 14. Branchfilters h_(C0)(v, f) to h_(C,K−1)(v, f) (1430, 1432, 1434) are generalnonlinearities with possible frequency dependence, as indicated by thetwo arguments v, the input signal, and f, the frequency. Inimplementation, they can take the form of monomial (cubic, quintic,etc.) memoryless nonlinearities. More general nonlinearities such asBessel functions or step functions, or any other convenientnonlinearity, may also be employed. One or more of these branch filtersmay instead have linear characteristics and frequency dependence. Forexample, they may take the form of delays or general linear filters, asin the aspect of the invention described immediately above. In the mostgeneral form, the branch filters depend on both the input signal andfrequency, where such dependencies may be intentional or inadvertent. Inthis model, the amplifier gain is included in the branch filterresponses. The branch filters 1430, 1432, and 1434 respectively precedeCGAs 1431, 1433, and 1435, the outputs of which are summed by combiner1436.

The filter H_(R)(f) 1410 in the reference branch may also be a simpledelay or a more general filter; even if such a filter is not insertedexplicitly, h_(r)(f) 1410 represents the response of the branch. Theobjective is to determine the responses of the observation filtersh_(p0)(f) to h_(p,K−1)(f) (1420, 1421, and 1422) at selectedfrequencies.

To determine the response h_(pk)(f_(i)) of the observation filter k atfrequency f_(i), the adaptation controller performs the followingactions:

(1) open the RF switch 1440, thereby disconnecting the filterH_(r)(f_(i)) 1410 from the subtractor 106;

(2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f_(i);

(3) set all CGA gains other than that for branch k to zero; select thebranch-k CGA gain to c′_(k) and the power of the input signal in someconvenient combination to cause the power amplifier to operate at apreselected output power that is common to all branches k andfrequencies f_(i) in this calibration procedure; doing so makes theamplifier gain and phase shift the same for all branches and frequenciesduring calibration;

(4) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal v_(e) with monitorsignal v_(emk)(f_(i)); the result is:

C _(ecmk)(f _(i))=c′ _(k) *h _(pk)*(f _(i))*P _(ck)(f _(i)),

where P_(ck)(f_(i)) is the power of signal v_(ck) at frequency f_(i);

(5) use a partial correlator, with local oscillators set to selectfrequency f_(i), to produce the correlation of signal monitorV_(cmk)(f_(i)) with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² *P _(ck)(f _(i));

(6) determine the branch-k observation filter response at frequencyf_(i) as:

h _(pk)(f _(i))=c′ _(k) C _(emk)(f _(i))/C _(ecmk)(f _(i)).

(7) close the RF switch.

The observation filter responses for the other filters are determinedsimilarly.

As will be apparent to those skilled in the art in light of theforegoing disclosure, many alterations and modifications are possible inthe practice of this invention without departing from the spirit orscope thereof. For example, the adjuster circuits of an analogpredistorter or a feedforward linearizer can employ both memory andnonlinearity in their branches. A cascade combination of a monomial anda filter within a branch is one way to accomplish this.

In addition, FIG. 5 illustrates a hybrid predistortion-feedforwardcircuit. The predistortion adjuster is implemented with a polynomialadjuster 109 as described above and shown in FIG. 2, and the distortionadjuster is implemented using the FIR adjuster 111 described above andshown in FIG. 4. The signal adjuster circuit 110 is a delay line havinga delay selected to match that of the polynomial adjuster 109 and thepower amplifier 103. The delay 112 of the delay line in the distortioncancellation circuit 102 is selected to match that of the FIR adjuster111 and error amplifier 108. The linearization of the power amplifiermay be improved by combining the predistortion and the feedforwardadjuster circuits. Further, rather than being a delay line, signaladjuster circuit 110 may be a FIR adjuster.

Other variations of the preceding linearizer circuits andself-calibration methods are deemed to be within the scope of thepresent invention, which is to be construed solely by the followingclaims.

What is claimed is:
 1. A linearizer comprising at least one observationfilter model and an adaptation controller, wherein the adaptationcontroller first determines a frequency response of said at least oneobservation filter model, and then corrects for the determined frequencyresponse.
 2. An amplifier comprising: a signal cancellation circuit; anda distortion cancellation circuit, wherein at least one of the signaland distortion cancellation circuits comprises at least one observationfilter model and an adaptation controller, and wherein the adaptationcontroller first determines a frequency response of said at least oneobservation filter model, and then corrects for the determined frequencyresponse.
 3. A method for an amplifier having at least one observationfilter model and an adaptation controller, said method comprising thesteps of: determining a frequency response of the at least oneobservation filter model; and correcting for the determined frequencyresponse.